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MAX17491GTA+ Datasheet(PDF) 9 Page - Maxim Integrated Products |
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MAX17491GTA+ Datasheet(HTML) 9 Page - Maxim Integrated Products |
9 / 12 page 5V Bias Supply (VDD) VDD provides the supply voltage for the internal logic cir- cuits. Bypass VDD with a 1µF or larger ceramic capaci- tor to GND to limit noise to the internal circuitry. Connect these bypass capacitors as close as possible to the IC. Input Undervoltage Lockout When VDD is below the UVLO threshold, DH and DL are held low. Once VDD is above the UVLO threshold and while PWM is low, DL is driven high and DH is driven low. This prevents the output of the converter from rising before a valid PWM signal is applied. Low-Power Pulse Skipping The MAX17491 enters into low-power pulse-skipping mode when SKIP is pulled low. In skip mode, an inherent automatic switchover to pulse-frequency modulation (PFM) takes place at light loads. A zero-crossing com- parator truncates the low-side switch on-time at the inductor current’s zero crossing. The comparator senses the voltage across LX and GND. Once VLX - VGND drops below the zero-crossing comparator threshold (see the Electrical Characteristics), the comparator forces DL low. This mechanism causes the threshold between pulse-skipping PFM and nonskipping PWM operation to coincide with the boundary between con- tinuous and discontinuous inductor-current operation. The PFM/PWM crossover occurs when the load current of each phase is equal to 1/2 the peak-to-peak ripple current, which is a function of the inductor value. For a battery input range of 7V to 20V, this threshold is rela- tively constant, with only a minor dependence on the input voltage due to the typically low duty cycles. The switching waveforms can appear noisy and asynchro- nous when light loading activates the pulse-skipping operation, but this is a normal operating condition that results in high light-load efficiency. Applications Information Power-MOSFET Selection Most of the following MOSFET guidelines focus on the challenge of obtaining high load-current capability when using high-voltage (> 20V) AC adapters. Low- current applications usually require less attention. The high-side MOSFET (NH) must be able to dissipate the resistive losses plus the switching losses at both VIN(MIN) and VIN(MAX). Calculate both these sums. Ideally, the losses at VIN(MIN) should be roughly equal to losses at VIN(MAX), with lower losses in between. If the losses at VIN(MIN) are significantly higher than the losses at VIN(MAX), consider increasing the size of NH (reducing RDS(ON) but increasing CGATE). Conversely, if the losses at VIN(MAX) are significantly higher than the losses at VIN(MIN), consider reducing the size of NH (increasing RDS(ON) but reducing CGATE). If VIN does not vary over a wide range, the minimum power dissi- pation occurs where the resistive losses equal the switching losses. Choose a low-side MOSFET that has the lowest possible on-resistance (RDS(ON)), comes in a moderate-sized package (i.e., one or two 8-pin SOs, DPAK, or D2PAK), and is reasonably priced. Ensure that the DL gate driver can supply sufficient current to support the gate charge and the current injected into the parasitic gate-to-drain capacitor caused by the high-side MOSFET turning on; otherwise, cross-con- duction problems can occur. MOSFET Power Dissipation Worst-case conduction losses occur at the duty factor extremes. For the high-side MOSFET (NH), the worst- case power dissipation due to resistance occurs at the minimum input voltage: where ηTOTAL is the total number of phases. Generally, a small high-side MOSFET is desired to reduce switch- ing losses at high input voltages. However, the RDS(ON) required to stay within package-power dissipation often limits how small the MOSFETs can be. Again, the opti- mum occurs when the switching losses equal the con- duction (RDS(ON)) losses. High-side switching losses do not usually become an issue until the input is greater than approximately 15V. Calculating the power dissipation in high-side MOSFETs (NH) due to switching losses is difficult since it must allow for difficult quantifying factors that influ- ence the turn-on and turn-off times. These factors include the internal gate resistance, gate charge, threshold voltage, source inductance, and PCB layout characteristics. The following switching-loss calculation provides only a very rough estimate and is no substitute for prototype evaluation, preferably including verification using a thermocouple mounted on NH: where COSS is the NH MOSFET’s output capacitance, QG(SW) is the charge needed to turn on the high-side MOSFET, and IGATE is the peak gate-drive source/sink current (5A typ). PD N SWITCHING VI f n Q I CV f H IN MAX LOAD SW TOTAL GSW GATE OSS IN SW ( ) () ( ) = ⎛ ⎝⎜ ⎞ ⎠⎟ ⎛ ⎝⎜ ⎞ ⎠⎟ + 2 2 PD N RESISTIVE V V I R H OUT IN LOAD TOTAL DS ON ( ) () = ⎛ ⎝⎜ ⎞ ⎠⎟ ⎛ ⎝⎜ ⎞ ⎠⎟ η 2 Single-Phase Synchronous MOSFET Driver _______________________________________________________________________________________ 9 |
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