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MC34151DG Datasheet(PDF) 7 Page - ON Semiconductor |
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MC34151DG Datasheet(HTML) 7 Page - ON Semiconductor |
7 / 12 page MC34151, MC33151 http://onsemi.com 7 the NPN pullup during the negative output transient, power dissipation at high frequencies can become excessive. Figures 20, 21, and 22 show a method of using external Schottky diode clamps to reduce driver power dissipation. Undervoltage Lockout An undervoltage lockout with hysteresis prevents erratic system operation at low supply voltages. The UVLO forces the Drive Outputs into a low state as VCC rises from 1.4 V to the 5.8 V upper threshold. The lower UVLO threshold is 5.3 V, yielding about 500 mV of hysteresis. Power Dissipation Circuit performance and long term reliability are enhanced with reduced die temperature. Die temperature increase is directly related to the power that the integrated circuit must dissipate and the total thermal resistance from the junction to ambient. The formula for calculating the junction temperature with the package in free air is: TJ =TA + PD (RqJA) where: TJ = Junction Temperature TA = Ambient Temperature PD = Power Dissipation RqJA = Thermal Resistance Junction to Ambient There are three basic components that make up total power to be dissipated when driving a capacitive load with respect to ground. They are: PD = PQ + PC + PT where: PQ = Quiescent Power Dissipation PC = Capacitive Load Power Dissipation PT = Transition Power Dissipation The quiescent power supply current depends on the supply voltage and duty cycle as shown in Figure 17. The device’s quiescent power dissipation is: PQ = VCC ICCL (1−D) + ICCH (D) where: ICCL = Supply Current with Low State Drive Outputs ICCH = Supply Current with High State Drive Outputs D = Output Duty Cycle The capacitive load power dissipation is directly related to the load capacitance value, frequency, and Drive Output voltage swing. The capacitive load power dissipation per driver is: PC =VCC (VOH − VOL) CL f where: VOH = High State Drive Output Voltage VOL = Low State Drive Output Voltage CL = Load Capacitance f = frequency When driving a MOSFET, the calculation of capacitive load power PC is somewhat complicated by the changing gate to source capacitance CGS as the device switches. To aid in this calculation, power MOSFET manufacturers provide gate charge information on their data sheets. Figure 18 shows a curve of gate voltage versus gate charge for the ON Semiconductor MTM15N50. Note that there are three distinct slopes to the curve representing different input capacitance values. To completely switch the MOSFET ‘on’, the gate must be brought to 10 V with respect to the source. The graph shows that a gate charge Qg of 110 nC is required when operating the MOSFET with a drain to source voltage VDS of 400 V. Qg, GATE CHARGE (nC) CGS = D Qg 16 12 8.0 4.0 0 0 40 80 120 160 VDS = 100 V VDS = 400 V 8.9 nF 2.0 nF MTM15N50 ID = 15 A TA = 25°C Figure 18. Gate−To−Source Voltage versus Gate Charge D VGS The capacitive load power dissipation is directly related to the required gate charge, and operating frequency. The capacitive load power dissipation per driver is: PC(MOSFET) = VC Qg f The flat region from 10 nC to 55 nC is caused by the drain−to−gate Miller capacitance, occurring while the MOSFET is in the linear region dissipating substantial amounts of power. The high output current capability of the MC34151 is able to quickly deliver the required gate charge for fast power efficient MOSFET switching. By operating the MC34151 at a higher VCC, additional charge can be provided to bring the gate above 10 V. This will reduce the ‘on’ resistance of the MOSFET at the expense of higher driver dissipation at a given operating frequency. The transition power dissipation is due to extremely short simultaneous conduction of internal circuit nodes when the Drive Outputs change state. The transition power dissipation per driver is approximately: PT = VCC (1.08 VCC CL f − 8 y 10−4) PT must be greater than zero. Switching time characterization of the MC34151 is performed with fixed capacitive loads. Figure 14 shows that for small capacitance loads, the switching speed is limited by transistor turn−on/off time and the slew rate of the internal nodes. For large capacitance loads, the switching speed is limited by the maximum output current capability of the integrated circuit. |
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