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LT1169CN8 Datasheet(PDF) 8 Page - Linear Technology |
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LT1169CN8 Datasheet(HTML) 8 Page - Linear Technology |
8 / 12 page 8 LT1169 S APPLICATI I FOR ATIO Figure 1. Comparison of LT1169, OP215, and AD822 Input Bias Current vs Common Mode Range SOURCE RESISTANCE ( Ω) 100 1 10 1k 10k 1k 100M 1G LT1169 • F02 100k 100 10M 10k 1M Vn = AV √Vn 2 (OP AMP) + 4kTR + 2qIBR 2 SOURCE RESISTANCE = 2RS = R * PLUS RESISTOR † PLUS RESISTOR 1000pF CAPACITOR RESISTOR NOISE ONLY LT1169 LT1124* LT1124† LT1169† LT1124 LT1169* CS CS RS RS VO Figure 2. Comparison of LT1169 and LT1124 Total Output 1kHz Voltage Noise vs Source Resistance the total noise. This means the LT1169 is superior to most dual JFET op amps. Only the lowest noise bipolar op amps have the advantage at low source resistances. As the source resistance increases from 5k to 50k, the LT1169 will match the best bipolar op amps for noise perfor- mance, since the thermal noise of the transducer (4kTR) begins to dominate the total noise. A further increase in source resistance, above 50k, is where the op amp’s current noise component (2qIBR2) will eventually domi- nate the total noise. At these high source resistances, the LT1169 will out perform the lowest noise bipolar op amps due to the inherently low current noise of FET input op amps. Clearly, the LT1169 will extend the range of high impedance transducers that can be used for high signal- to-noise ratios. This makes the LT1169 the best choice for high impedance, capacitive transducers. Optimization Techniques for Charge Amplifiers The high input impedance JFET front end makes the LT1169 suitable in applications where very high charge sensitivity is required. Figure 3 illustrates the LT1169 in its inverting and noninverting modes of operation. A charge amplifier is shown in the inverting mode example; the gain depends on the principal of charge conservation at the input of the LT1169. The charge across the transducer capacitance CS is transferred to the feedback capacitor CF resulting in a change in voltage dV, which is equal to dQ/CF. The gain therefore is 1 + CF/CS. For unity-gain, the CF should equal the transducer capacitance plus the input capacitance of the LT1169 and RF should equal RS. In the noninverting mode example, the transducer current is converted to a change in voltage by the transducer capacitance, CS. This voltage is then buffered by the LT1169 with a gain of 1 + R1/R2. A DC path is provided by RS, which is either the transducer impedance or an external resistor. Since RS is usually several orders of magnitude greater than the parallel combination of R1 and R2, RB is added to balance the DC offset caused by the noninverting input bias current and RS. The input bias currents, although small at room temperature, can create significant errors over increasing temperature, especially with transducer resistances of up to 1000M Ω or more. The optimum value for RB is determined by equating the thermal noise (4kTRS) to the current noise (2qIB) times RS2. Solving for RS results in RB = RS = 2VT/IB. A parallel Amplifying Signals from High Impedance Transducers The low voltage and current noise offered by the LT1169 makes it useful in a wide range of applications, especially where high impedance, capacitive transducers are used such as hydrophones, precision accelerometers, and photodiodes. The total output noise in such a system is the gain times the RMS sum of the op amp’s input referred voltage noise, the thermal noise of the transducer, and the op amp’s input bias current noise times the transducer impedance. Figure 2 shows total input voltage noise versus source resistance. In a low source resistance (< 5k) application the op amp voltage noise will dominate COMMON MODE RANGE (V) –15 –100 –60 –40 –20 0 20 40 –10 –5 05 LT1169 • F01 10 60 80 100 –80 15 LT1169 AD822 CURRENT NOISE = √2qIB OP215 |
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