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OPA643N-250 Datasheet(PDF) 11 Page - Texas Instruments |
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OPA643N-250 Datasheet(HTML) 11 Page - Texas Instruments |
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11 / 18 page  ® OPA643 11 This will increase the Q for the closed-loop poles, peaking up the frequency response and extending the bandwidth. A peaked frequency response will show overshoot and ringing in the pulse response as well as a higher integrated output noise. Operating at a noise gain less than +3 runs the risk of sustained oscillation (loop instability). However, operation at low gains would be desirable to take advantage of the much higher slew rate and lower input noise voltage available in the OPA643, as compared to performance offered by unity gain stable op amps. Numerous external compensation techniques have been suggested for operating a high gain op amp at low gains. Most of these give zero/pole pairs in the closed-loop response that cause long term settling tails in the pulse response and/or phase non-linearity in the frequency response. Figure 5 shows an external compensation method for the non-inverting configuration that does not suffer from these drawbacks. gain for the op amp and the noise gain pole, set by 1/RFCF, is placed correctly, a very well controlled second-order low pass frequency response will result. R F 402 Ω R I 133 Ω R G 402 Ω R T 50 Ω 50 Ω OPA643 +5V –5V V O 50 Ω Source FIGURE 5. Broadband Low Gain Non-Inverting External Compensation. The RI resistor across the two inputs will increase the noise gain (i.e. decrease the loop gain) without changing the signal gain. This approach will retain the full slew rate to the output but will give up some of the low noise benefit of the OPA643. Assuming a low source impedance, set RI so that 1+R F/(RG || RI) is ≥ +3. Where a low gain is desired, and inverting operation is acceptable, a new external compensation technique may be used to retain the full slew rate and noise benefits of the OPA643 while maintaining the increased loop gain and the associated improvement in distortion offered by the decompensated architecture. This technique shapes the loop gain for good stability while giving an easily controlled second-order low pass frequency response. Figure 6 shows this circuit (the same amplifier circuit as shown on the front page). Considering only the noise gain for the circuit of Figure 6, the low frequency noise gain, (NG1) will be set by the resistor ratios while the high frequency noise gain (NG2) will be set by the capacitor ratios. The capacitor values set both the transition frequencies and the high frequency noise gain. If this noise gain, determined by NG2 = 1+ CS/CF, is set to a value greater than the recommended minimum stable R F 806 Ω C S 12.6pF 0.1µF OPA643 +5V –5V V O V I C F 1.9pF R G 402 Ω R T 280 Ω FIGURE 6. Broadband Low Gain Inverting External Compensation. To choose the values for both CS and CF, two parameters and only three equations need to be solved. The first parameter is the target high frequency noise gain NG2, which should be greater than the minimum stable gain for the OPA643. Here, a target NG2 of 7.5 will be used. The second parameter is the desired low frequency signal gain, which also sets the low frequency noise gain NG1. To simplify this discussion, we will target a maximally flat second-order low pass Butterworth frequency response (Q = 0.707). The signal gain of –2 shown in Figure 6 will set the low frequency noise gain to NG1 = 1 + RF/RG (= 3 in this example). Then, using only these two gains and the Gain Bandwidth Product (GBP) for the OPA643 (800MHz), the key frequency in the compensation can be determined as: Physically, this Z 0 (13.6MHz for the values shown in Figure 6) is set by 1/(2 π • R F(CF + CS)) and is the frequency at which the rising portion of the noise gain would intersect unity gain if projected back to 0dB gain. The actual zero in the noise gain occurs at NG 1 • Z0 and the pole in the noise gain occurs at NG2 • Z0. Since GBP is expressed in Hz, multiply Z 0 by 2π and use this to get CF by solving: Finally, since CS and CF set the high frequency noise gain, determine CS by: The resulting closed-loop bandwidth will be approximately equal to: F –3dB ≅ Z O GBP Z O = GBP NG 1 2 1– NG 1 NG 2 –1 – 2 NG 1 NG 2 C F = 1 2 π• R F Z O NG2 C S = NG2 –1 () C F |
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