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LT8331 Datasheet(HTML) 17 Page - Analog Devices

Part No. LT8331
Description  Low IQ Boost/SEPIC/Flyback/Inverting Converter with 0.5A, 140V Switch
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Homepage  http://www.analog.com
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LT8331 Datasheet(HTML) 17 Page - Analog Devices

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LT8331
17
Rev. C
For more information www.analog.com
APPLICATIONS INFORMATION
the converter performance. A higher duty cycle affects the
flyback converter in the following aspects:
n
Lower switch RMS current ISW(RMS), but higher switch
VSW peak voltage
n
Lower diode peak reverse voltage, but higher diode
RMS current ID(RMS)
n
Higher transformer turns ratio (NP/NS)
It is recommended to choose a duty cycle between 20%
and 80%.
Flyback Converter: Maximum Output Current
Capability and Transformer Design
The maximum output current capability and transformer
design for continuous conduction mode (CCM) is chosen
as presented here.
The maximum duty cycle (DMAX) occurs when the con-
verter has the minimum VIN:
DMAX =
VOUT
NP
NS


VOUT
NP
NS

 + VIN(MIN)
Due to the current limit of its internal power switch, the
LT8331 should be used in a flyback converter whose
maximum output current (IO(MAX)) is:
IO(MAX)
VIN(MIN)
VOUT
• DMAX • 0.5A − 0.5 • ΔISW
(
) • η
where η (< 1.0) is the converter efficiency. Minimum
possible inductor value and switching frequency should
also be considered since they will increase inductor ripple
current ∆ISW.
The transformer ripple current ∆ISW has a direct effect on
the design/choice of the transformer and the converter’s
output current capability. Choosing smaller values of ∆ISW
increases the output current capability, but requires large
primary and secondary inductances and reduces the cur-
rent loop gain (the converter will approach voltage mode).
Accepting larger values of ∆ISW allows the use of low
primary and secondary inductances, but results in higher
input current ripple, greater core losses, and reduces the
output current capability. It is recommended to choose a
∆ISW of approximately 0.2A to 0.3A.
Given an operating input voltage range, and having cho-
sen the operating frequency and ripple current in the pri-
mary winding, the primary winding inductance can be
calculated using the following equation:
L =
VIN(MIN)
ΔISW • fOSC
• DMAX
The primary winding peak current is the switch current
limit (maximum 0.7A). The primary and secondary maxi-
mum RMS currents are:
ILP(RMS)
POUT(MAX)
DMAX • VIN(MIN) • η
ILS(RMS)
IOUT(MAX)
1 − DMAX
Based on the preceding equations, the user should design/
choose the transformer having sufficient saturation and
RMS current ratings.
Flyback Converter: Snubber Design
Transformer leakage inductance (on either the primary
or secondary) causes a voltage spike to occur after the
MOSFET turn-off. This is increasingly prominent at higher
load currents, where more stored energy must be dis-
sipated. In some cases a snubber circuit will be required
to avoid overvoltage breakdown at the MOSFET’s drain
node. There are different snubber circuits (such as RC
snubber, RCD snubber, etc.) and Application Note 19 is
a good reference on snubber design. An RCD snubber is
shown in Figure 6.
The snubber resistor value (RSN) can be calculated by the
following equation:
RSN = 2 •
V2SN − VSN • VOUT
NP
NS
I2SW(PEAK) • LLK • fOSC


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