Electronic Components Datasheet Search |
|
74HCT9046AD Datasheet(PDF) 9 Page - NXP Semiconductors |
|
74HCT9046AD Datasheet(HTML) 9 Page - NXP Semiconductors |
9 / 40 page 1999 Jan 11 9 Philips Semiconductors Product specification PLL with bandgap controlled VCO 74HCT9046A The pump current IP is independent from the supply voltage and is set by the internal bandgap reference of 2.5 V. Rb is the external bias resistor between pin 15 and ground. The current and voltage transfer function of PC2 are shown in Fig.9. The phase comparator gain is: Typical waveforms for the PC2 loop locked at fc are shown in Fig.10. When the frequencies of SIGIN and COMPIN are equal but the phase of SIGIN leads that of COMPIN, the up output driver at PC2OUT is held ‘ON’ for a time corresponding to the phase difference ( ΦPCIN). When the phase of SIGIN lags that of COMPIN, the down or sink driver is held ‘ON’. When the frequency of SIGIN is higher than that of COMPIN, the source output driver is held ‘ON’ for most of the input signal cycle time and for the remainder of the cycle time both drivers are ‘OFF’ (3-state). If the SIGIN frequency is lower than the COMPIN frequency, then it is the sink driver that is held ‘ON’ for most of the cycle. Subsequently the voltage at the capacitor (C2) of the low-pass filter connected to PC2OUT varies until the signal and comparator inputs are equal in both phase and frequency. At this stable point the voltage on C2 remains constant as the PC2 output is in 3-state and the VCO input at pin 9 is a high impedance. Also in this condition the signal at the phase comparator pulse output (PCPOUT) has a minimum output pulse width equal to the overlap time, so can be used for indicating a locked condition. I P 17 2.5 R b -------- A () × = K p I P 2 π ------- Ar ⁄ () = Thus for PC2 no phase difference exists between SIGIN and COMPIN over the full frequency range of the VCO. Moreover, the power dissipation due to the low-pass filter is reduced because both output drivers are OFF for most of the signal input cycle. It should be noted that the PLL lock range for this type of phase comparator is equal to the capture range and is independent of the low-pass filter. With no signal present at SIGIN the VCO adjust, via PC2, to its lowest frequency. By using current sources as charge pump output on PC2, the dead zone or backlash time could be reduced to zero. Also, the pulse widening due to the parasitic output capacitance plays no role here. This enables a linear transfer function, even in the vicinity of the zero crossing. The differences between a voltage switch charge pump and a current switch charge pump are shown in Fig.11. The design of the low-pass filter is somewhat different when using current sources. The external resistor R3 is no longer present when using PC2 as phase comparator. The current source is set by Rb. A simple capacitor behaves as an ideal integrator now, because the capacitor is charged by a constant current. The transfer function of the voltage switch charge pump may be used. In fact it is even more valid, because the transfer function is no longer restricted for small changes only. Further the current is independent from both the supply voltage and the voltage across the filter. For one that is familiar with the low-pass filter design of the 4046A a relation may show how Rb relates with a fictive series resistance, called R3'. This relation can be derived by assuming first that a voltage controlled switch PC2 of the 4046A is connected to the filter capacitance C2 via this fictive R3' (see Fig.8b). Then during the PC2 output pulse the charge current equals: With the initial voltage VC2(0) at: 1 ⁄2VCC = 2.5 V, As shown before the charge current of the current switch of the 9046A is: Hence: Using this equivalent resistance R3' for the filter design the voltage can now be expressed as a transfer function of PC2; assuming ripple (fr =fi) is suppressed, as: Again this illustrates the supply voltage independent behaviour of PC2. Examples of PC2 combined with a passive filter are shown in Figs 12 and 13. Figure 12 shows that PC2 with only a C2 filter behaves as a high-gain filter. For stability the damped version of Fig.13 with series resistance R4 is preferred. Practical design values for Rb are between 25 and 250 k Ω with R3' = 1.5 to 15 k Ω for the filter design. Higher values for R3' require lower values for the filter capacitance which is very advantageous at low values the loop natural frequency ω n. I P V CC V C2 0 () – R3' ----------------------------------- = I P 2.5 R3' --------- = I P 17 2.5 R b -------- × = R3' R b 17 ------- Ω () = K PC2 5 4 π ------- Vr ⁄ () = |
Similar Part No. - 74HCT9046AD |
|
Similar Description - 74HCT9046AD |
|
|
Link URL |
Privacy Policy |
ALLDATASHEET.NET |
Does ALLDATASHEET help your business so far? [ DONATE ] |
About Alldatasheet | Advertisement | Contact us | Privacy Policy | Link Exchange | Manufacturer List All Rights Reserved©Alldatasheet.com |
Russian : Alldatasheetru.com | Korean : Alldatasheet.co.kr | Spanish : Alldatasheet.es | French : Alldatasheet.fr | Italian : Alldatasheetit.com Portuguese : Alldatasheetpt.com | Polish : Alldatasheet.pl | Vietnamese : Alldatasheet.vn Indian : Alldatasheet.in | Mexican : Alldatasheet.com.mx | British : Alldatasheet.co.uk | New Zealand : Alldatasheet.co.nz |
Family Site : ic2ic.com |
icmetro.com |