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OPA847IDG4 Datasheet(PDF) 11 Page - Texas Instruments |
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OPA847IDG4 Datasheet(HTML) 11 Page - Texas Instruments |
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11 / 30 page  OPA847 11 SBOS251E www.ti.com WIDEBAND, HIGH SENSITIVITY, TRANSIMPEDANCE DESIGN The high GBP and low input voltage and current noise for the OPA847 make it an ideal wideband transimpedance ampli- fier for low to moderate transimpedance gains. Very high transimpedance gains (> 100k Ω) will benefit from the low input noise current of a JFET input op amp such as the OPA657. Unity-gain stability in the op amp is not required for application as a transimpedance amplifier. Figure 3 shows one possible transimpedance design example that would be particularly suitable for the 155Mbit data rate of an OC-3 receiver. Designs that require high bandwidth from a large area detector with relatively low transimpedance gain will benefit from the low input voltage noise for the OPA847. The amplifier’s input voltage noise is peaked up over frequency by the diode source capacitance, and can (in many cases) become the limiting factor to input sensitivity. The key ele- ments to the design are the expected diode capacitance (CD) with the reverse bias voltage (–VB) applied, the desired transimpedance gain (RF), and the GBP for the OPA847 (3900MHz). With these three variables set (including the parasitic input capacitance for the OPA847 added to CD), the feedback capacitor value (CF) can be set to control the frequency response. Equation 2 gives the approximate –3dB bandwidth that results if CF is set using Equation 1. f GBP RC Hz dB FD − = () 3 2 π (2) The example of Figure 3 gives approximately 104MHz flat bandwidth using the 0.18pF feedback compensation capaci- tor. This bandwidth easily supports an OC-3 receiver with exceptional sensitivity. If the total output noise is bandlimited to a frequency less than the feedback pole frequency, a very simple expression for the equivalent input noise current is shown as Equation 3. (3) ii kT R EC F EQ N F ND =+ + ( ) 2 2 2 4 2 3 π where: iEQ = Equivalent input noise current if the output noise is bandlimited to f < 1/2 πR FCF iN = Input current noise for the op amp inverting input eN = Input voltage noise for the op amp CD = Total Inverting Node Capacitance f = Bandlimiting frequency in Hz (usually a post filter prior to further signal processing) Evaluating this expression up to the feedback pole frequency at 74MHz for the circuit of Figure 3 gives an equivalent input noise current of 3.0pA/ √Hz. This is slightly higher than the 2.5pA/ √Hz input current noise for the op amp. This total equivalent input current noise is slightly increased by the last term in the equivalent input noise expression. It is essential in this case to use a low-voltage noise op amp. For example, if a slightly higher input noise voltage, but otherwise identical, op amp were used instead of the OPA847 in this application (say 2.0nV/ √Hz), the total input referred current noise would increase to 3.7pA/ √Hz. Low input voltage noise is required for the best sensitivity in these wideband transimpedance applications. This is often unspecified for dedicated transim- pedance amplifiers with a total output noise for a specified source capacitance given instead. It is the relatively high input voltage noise for those components that cause higher than expected output noise if the source capacitance is higher than specified. The output DC error for the circuit of Figure 3 is minimized by including a 12k Ω to ground on the noninverting input. This reduces the contribution of input bias current errors (for total output offset voltage) to the offset current times the feedback resistor. To minimize the output noise contribution of this resistor, 0.01 µF and 100pF capacitors are included in paral- lel. Worst-case output DC error for the circuit of Figure 3 at 25 °C is: VOS = ±0.5mV (input offset voltage) ± 0.6µA (input offset current) • 12k Ω = ±7.2mV Worst-case output offset DC drift (over the 0 °C to 70°C span) is: dVOS/dT = ±1.5µV/°C (input offset drift) ± 2nA/°C (input offset current drift) • 12k Ω = ±21.5µV/°C. To achieve a maximally flat 2nd-order Butterworth frequency response, set the feedback pole as shown in Equation 1. 1 24 ππ RC GBP RC FF FD = (1) Adding the common-mode and differential mode input ca- pacitance (1.2 + 2.5)pF to the 1pF diode source capacitance of Figure 3, and targeting a 12k Ω transimpedance gain using the 3900MHz GBP for the OPA847 requires a feedback pole set to 74MHz to get a nominal Butterworth frequency re- sponse design. This requires a total feedback capacitance of 0.18pF. That total is shown in Figure 3, but recall that typical surface-mount resistors have a parasitic capacitance of 0.2pF, leaving no external capacitor required for this design. FIGURE 3. Wideband, High Sensitivity, OC-3 Transimpedance Amplifier. R F 12k Ω 12k Ω 0.1 µF 100pF Power-supply decoupling not shown. λ OPA847 +5V –5V –V B C F 0.18pF 1pF Photodiode V DIS |
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