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HIP6004 Datasheet(PDF) 8 Page - Intersil Corporation |
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HIP6004 Datasheet(HTML) 8 Page - Intersil Corporation |
8 / 12 page 8 The modulator transfer function is the small-signal transfer function of VOUT/VE/A. This function is dominated by a DC Gain and the output filter (LO and CO), with a double pole break frequency at FLC and a zero at FESR. The DC Gain of the modulator is simply the input voltage (VIN) divided by the peak-to-peak oscillator voltage ∆VOSC. Modulator Break Frequency Equations The compensation network consists of the error amplifier (internal to the HIP6004) and the impedance networks ZIN and ZFB. The goal of the compensation network is to provide a closed loop transfer function with the highest 0dB crossing frequency (f0dB) and adequate phase margin. Phase margin is the difference between the closed loop phase at f0dB and 180 degrees . The equations below relate the compensation network’s poles, zeros and gain to the components (R1, R2, R3, C1, C2, and C3) in Figure 8. Use these guidelines for locating the poles and zeros of the compensation network: 1. Pick Gain (R2/R1) for desired converter bandwidth 2. Place 1STZero Below Filter’s Double Pole (~75% FLC) 3. Place 2ND Zero at Filter’s Double Pole 4. Place 1ST Pole at the ESR Zero 5. Place 2ND Pole at Half the Switching Frequency 6. Check Gain against Error Amplifier’s Open-Loop Gain 7. Estimate Phase Margin - Repeat if Necessary Compensation Break Frequency Equations Figure 9 shows an asymptotic plot of the DC-DC converter’s gain vs. frequency. The actual Modulator Gain has a high gain peak due to the high Q factor of the output filter and is not shown in Figure 9. Using the above guidelines should give a Compensation Gain similar to the curve plotted. The open loop error amplifier gain bounds the compensation gain. Check the compensation gain at FP2 with the capabilities of the error amplifier. The Closed Loop Gain is constructed on the log-log graph of Figure 9 by adding the Modulator Gain (in dB) to the Compensation Gain (in dB). This is equivalent to multiplying the modulator transfer function to the compensation transfer function and plotting the gain. The compensation gain uses external impedance networks ZFB and ZIN to provide a stable, high bandwidth (BW) overall loop. A stable control loop has a gain crossing with -20dB/decade slope and a phase margin greater than 45 degrees. Include worst case component variations when determining phase margin. Component Selection Guidelines Output Capacitor Selection An output capacitor is required to filter the output and supply the load transient current. The filtering requirements are a function of the switching frequency and the ripple current. The load transient requirements are a function of the slew rate (di/dt) and the magnitude of the transient load current. These requirements are generally met with a mix of capacitors and careful layout. Modern microprocessors produce transient load rates above 1A/ns. High frequency capacitors initially supply the transient and slow the current load rate seen by the bulk capacitors. The bulk filter capacitor values are generally determined by the ESR (effective series resistance) and voltage rating requirements rather than actual capacitance requirements. High frequency decoupling capacitors should be placed as close to the power pins of the load as physically possible. Be careful not to add inductance in the circuit board wiring that could cancel the usefulness of these low inductance components. Consult with the manufacturer of the load on specific decoupling requirements. For example, Intel recommends that the high frequency decoupling for the Pentium Pro be composed of at least forty (40) 1 µF ceramic capacitors in the 1206 surface-mount package. Use only specialized low-ESR capacitors intended for switching-regulator applications for the bulk capacitors. The bulk capacitor’s ESR will determine the output ripple voltage and the initial voltage drop after a high slew-rate transient. An aluminum electrolytic capacitor’s ESR value is related to the case size with lower ESR available in larger case sizes. However, the equivalent series inductance (ESL) of these capacitors increases with case size and can reduce the usefulness of the capacitor to high slew-rate transient loading. Unfortunately, ESL is not a specified parameter. Work with your capacitor supplier and measure the capacitor’s impedance with frequency to select a suitable component. In F LC 1 2 π L O C O • • --------------------------------------- = F ESR 1 2 π ESR C O • • ---------------------------------------- = F Z1 1 2 π R2 • C1 • ---------------------------------- = F Z2 2 π R1 R3 + () C3 • • = F P1 1 2 π R 2 C1 C2 • C1 C2 + ---------------------- • • ------------------------------------------------------ = F P2 1 2 π R3 • C3 • ---------------------------------- = 100 80 60 40 20 0 -20 -40 -60 FP1 FZ2 10M 1M 100K 10K 1K 100 10 OPEN LOOP ERROR AMP GAIN FZ1 FP2 20LOG FLC FESR COMPENSATION FREQUENCY (Hz) GAIN 20LOG (VIN/∆VOSC) MODULATOR GAIN (R2/R1) FIGURE 9. ASYMPTOTIC BODE PLOT OF CONVERTER GAIN CLOSED LOOP GAIN HIP6004 |
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