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LT1676I Datasheet(PDF) 8 Page - Linear Technology |
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LT1676I Datasheet(HTML) 8 Page - Linear Technology |
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8 / 16 page  8 LT1676 APPLICATIONS INFORMATION Once the inductance value is decided, inductor peak current rating and resistance need to be considered. Here, the inductor peak current rating refers to the onset of saturation in the core material, although manufacturers sometimes specify a “peak current rating” which is derived from a worst-case combination of core saturation and self-heating effects. Inductor winding resistance alone limits the inductor’s current carrying capability as the I2R power threatens to overheat the inductor. If applicable, remember to include the condition of output short circuit. Although the peak current rating of the inductor can be exceeded in short-circuit operation, as core saturation per se is not destructive to the core, excess resistive self- heating is still a potential problem. The final inductor selection is generally based on cost, which usually translates into choosing the smallest physi- cal size part that meets the desired inductance value, resistance and current carrying capability. An additional factor to consider is that of physical construction. Briefly stated, “open” inductors built on a rod- or barrel-shaped core generally offer the smallest physical size and lowest cost. However their open construction does not contain the resulting magnetic field, and they may not be accept- able in RFI-sensitive applications. Toroidal style induc- tors, many available in surface mount configuration, offer improved RFI performance, generally at an increase in cost and physical size. And although custom design is always a possibility, most potential LT1676 applications can be handled by the array of standard, off-the-shelf inductor products offered by the major suppliers. Selecting Freewheeling Diode Highest efficiency operation requires the use of a Schottky type diode. DC switching losses are minimized due to its low forward voltage drop, and AC behavior is benign due to its lack of a significant reverse recovery time. Schottky diodes are generally available with reverse voltage ratings of 60V and even 100V, and are price competitive with other types. The use of so-called “ultrafast” recovery diodes is gener- ally not recommended. When operating in continuous mode, the reverse recovery time exhibited by “ultrafast” diodes will result in a slingshot type effect. The power internal switch will ramp up VIN current into the diode in an attempt to get it to recover. Then, when the diode has finally turned off, some tens of nanoseconds later, the VSW node voltage ramps up at an extremely high dV/dt, per- haps 5 to even 10V/ns ! With real world lead inductances, the VSW node can easily overshoot the VIN rail. This can result in poor RFI behavior and if the overshoot is severe enough, damage the IC itself. Selecting Bypass Capacitors The basic topology as shown in Figure 1 uses two bypass capacitors, one for the VIN input supply and one for the VOUT output supply. User selection of an appropriate output capacitor is rela- tively easy, as this capacitor sees only the AC ripple current in the inductor. As the LT1676 is designed for Buck or step-down applications, output voltage will nearly always be compatible with tantalum type capacitors, which are generally available in ratings up to 35V or so. These tantalum types offer good volumetric efficiency and many are available with specified ESR performance. The product of inductor AC ripple current and output capacitor ESR will manifest itself as peak-to-peak voltage ripple on the output node. (Note: If this ripple becomes too large, heavier control loop compensation, at least at the switching fre- quency, may be required on the VC pin.) The most demanding applications, requiring very low output ripple, may be best served not with a single extremely large output capacitor, but instead by the common technique of a separate L/C lowpass post filter in series with the output. (In this case, “Two caps are better than one.”) The input bypass capacitor is normally a more difficult choice. In a typical application e.g., 48VIN to 5VOUT, relatively heavy VIN current is drawn by the power switch for only a small portion of the oscillator period (low ON duty cycle). The resulting RMS ripple current, for which the capacitor must be rated, is often several times the DC average VIN current. Similarly, the “glitch” seen on the VIN supply as the power switch turns on and off will be related to the product of capacitor ESR, and the relatively high instantaneous current drawn by the switch. To compound these problems is the fact that most of these applications will be designed for a relatively high input voltage, for |
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