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ADP3164RUZ-R71 Datasheet(PDF) 11 Page - ON Semiconductor |
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ADP3164RUZ-R71 Datasheet(HTML) 11 Page - ON Semiconductor |
11 / 15 page REV. 0 ADP3164 –11– COUT Selection The required equivalent series resistance (ESR) and capacitance drive the selection of the type and quantity of the output capaci- tors. The ESR must be less than or equal to the specified output resistance (ROUT), in this case 0.95 m . The capacitance must be large enough that the voltage across the capacitors, which is the sum of the resistive and capacitive voltage deviations, does not deviate beyond the initial resistive step while the inductor current ramps up or down to the value corresponding to the new load current. One can, for example, use thirteen SP-Type OS-CON capaci- tors from Sanyo, with 820 F capacitance, a 4 V voltage rating, and 12 m ESR. The ten capacitors have a maximum total ESR of 0.92 m when connected in parallel. As long as the capacitance of the output capacitor bank is above a critical value and the regulating loop is compensated with Analog Devices’ proprietary compensation technique (ADOPT), the actual capacitance value has no influence on the peak-to- peak deviation of the output voltage to a full step change in the load current. The critical capacitance can be calculated as follows: C I RV L n A mV nH mF OUT CRIT O OUT OUT () .. . 80 0 95 1 475 600 4 856 (13) The critical capacitance limit for this circuit is 8.56 mF, while the actual capacitance of the thirteen OS-CON capacitors is 13 820 F = 10.66 mF. In this case, the capacitance is safely above the critical value. Multilayer ceramic capacitors are also required for high-frequency decoupling of the processor. The exact number of these MLC capacitors is a function of the board layout space and parasitics. Typical designs use twenty to thirty 10 F MLC capacitors located as close to the processor power pins as is practical. Feedback Loop Compensation Design for ADOPT Optimized compensation of the ADP3164 allows the best pos- sible containment of the peak-to-peak output voltage deviation. Any practical switching power converter is inherently limited by the inductor in its output current slew rate to a value much less than the slew rate of the load. Therefore, any sudden change of load current will initially flow through the output capacitors, and assuming that the capacitance of the output capacitor is larger than the critical value defined by Equation 13, this will produce a peak output voltage deviation equal to the ESR of the output capacitor times the load current change. The optimal implementation of voltage positioning, ADOPT, will create an output impedance of the power converter that is entirely resistive over the widest possible frequency range, includ- ing dc, and equal to the maximum acceptable ESR of the output capacitor array. With the resistive output impedance, the output voltage will droop in proportion with the load current at any load current slew rate; this ensures the optimal positioning and allows the minimization of the output capacitor bank. With an ideal current-mode-controlled converter, where the average inductor current would respond without delay to the command signal, the resistive output impedance could be achieved by having a single-pole roll-off of the voltage gain of the voltage-error amplifier. The pole frequency must coincide with the ESR zero of the output capacitor bank. The ADP3164 uses constant frequency current-mode control, which is known to have a nonideal, frequency-dependent command signal to inductor current transfer function. The frequency dependence manifests in the form of a pair of complex conjugate poles at one-half of the switching frequency. A purely resistive output impedance could be achieved by canceling the complex conjugate poles with zeros at the same complex frequencies and adding a third pole equal to the ESR zero of the output capacitor. Such compensating network would be quite complicated. Fortunately, i practice it is sufficient to cancel the pair of complex conjugate poles with a single real zero placed at one-half of the switching frequency. Although the end result is not a perfectly resistive output impedance, the remaining frequency dependence causes only a small percentage of deviation from the ideal resistive response. The single-pole and single-zero compensation can easily be implemented by terminating the gm error amplifier with the parallel combination of a resistor (RT) and a series RC network The value of the terminating resistor RT was previously deter- mined; the capacitance and resistance of the series RC network are calculated as follows: C CR R n fR C mF m k kHz k nF OC OUT OUT T OSC T OC 10 7 0 92 748 4 800 7 48 11 .. .. . (14 The nearest standard value of COC is 1 nF. The resistance of th zero-setting resistor in series with the compensating capacitor is R n f C kHz nF k Z OSC OC 4 800 1 159 . (15 The nearest standard 5% resistor value is 1.5 k . Note that this resistor is only required when COUT approaches CCRIT (within 25% or less). In this example, COUT is approaching CCRIT, so RZ should be included. Power MOSFETs In this example, eight N-channel power MOSFETs must be used; four as the main (control) switches, and the remaining four as the synchronous rectifier switches. The main selection parameters for the power MOSFETs are VGS(TH), QG and RDS(ON). The minimum gate drive voltage (the supply voltage to the ADP3414) dictates whether standard threshold or logic-level threshold MOSFETs must be used. Since VGATE <8 V, logic-level thresh- old MOSFETs (VGS(TH) < 2.5 V) are strongly recommended. Rev. 1 | Page 11 of 15 | www.onsemi.com Rev. 2 | Page 11 of 15 | www.onsemi.com |
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